Such a radar device is known from the patent publication DE 100 50 278 B4. It can be used to determine, via a LFMSK-transmission method, the distance and/or relative speed of a vehicle in reference to an object. The abbreviation here represents Linear Frequency Modulated Shift Keying.
A radar device with the above-mentioned features of the applicant is explained in greater detail based on prior art FIGS. 1 and 2.
FIG. 1 shows the control means 1, the oscillator 2, the transmission antenna 3, the receiving antenna 4, and the mixer 5.
The control means can address the oscillator 2. For this purpose, an output of the control means 1 is connected to a digital-analog converter 9, which converts a predetermined digital value into an analog voltage.
By addressing the oscillator 2 via the control means 1 said oscillator 2 can generate the signal such that it shows different signal portions A, B, C, . . . . Each signal portion A, B, C, . . . of the signal shows a sequence of signal fragments Ai, Bi, Ci, . . . . First signal fragments A1, B1, C1, . . . of the various signal portions A, B, C, . . . of the signal show different frequencies f1A, f1B, f1C, . . . and follow each other. The first signal fragments A1, B1, C1, . . . are followed by additional signal fragments Ai, Bi, Ci, . . . of the signal portion A, B, C, . . . of the signal, which also follow each other.
The frequency fiA, fiB, fiC, . . . of the signal fragments Ai, Bi, Ci, . . . of a signal portion A, B, C, . . . of the signals increases during one cycle either from one signal fragment to the next signal fragment by one frequency increment Δf or decreases during the cycle from one signal fragment to the next signal fragment by the frequency increment Δf. It is also possible for the frequency fiA, fiB , fiC, . . . of the signal fragments Ai, Bi, Ci, . . . of a signal portion A, B, C, . . . to remain steady during a cycle.
The signal fragments, also called bursts, show for example a length of 25 μs. The frequency fiA, fiB, fiC, . . . within a signal fragment Ai, Bi, Ci, . . . of an arbitrary signal portion A, B, C, . . . of the signal is constant and during a cycle, based on a frequency f0A, f0B, f0C, . . . of a signal fragment A0, B0, C0, . . . of a single portion A, B, C, . . . it can be increased or reduced to the next signal fragment A1, B1, C1, . . . of the same signal portion A, B, C, . . . of the signal by a preferably fixed frequency increment Δf or remain steady. Therefore, e. g. it results for the signal portion A: fiA=f0A+i·Δf with i=1, . . . , N−1. A typical value for N, i.e. for the number of signal fragments Ai, Bi, Ci, . . . of a signal portion A, B, C, . . . per cycle is 512. Depending on the frequency of the signal fragments within a cycle being increased or reduced, i.e. represents, Δf>0, Δf<0 or Δf=0, a cycle is also called upchirp, downchirp, or Doppler chirp. Upchirps, downchirps, and/or Doppler chirps are preferably transmitted alternating.
The different signal portions A, B, C, . . . of the signal are nested, i.e. here the signal fragments Ai, Bi, Ci, . . . of the various signal portions A, B, C, . . . follow each other in a preferably fixed sequence, as shown in FIG. 2 for an upchirp . The frequencies of the first signal fragments A, B, C, . . . of the various signal portions A, B, C, . . . are distinguished from each other by a difference which is very small in reference to the frequencies. The frequency increments Δf are also very small in reference to the frequencies of the various signal portions A, B, C, . . . , when they are above or below zero.
The frequency range of a cycle of 38.4 ms, respectively swept through by the signals A, B, or C, typically amounts to 90 MHz. The frequency differences fiB−fiA and/or fiC−fiB each amount to approximately 1.2 MHz. The selection of these parameters, in addition to statutory stipulations of the covered band width of the allocated frequency band, is primarily determined by the requirements of the target detection, which shall occur by the radar device.
The signal generated by the oscillator 2 is transmitted by the transmitting antenna 3.
The signal transmitted by the transmitting antenna 3 may be reflected by one or more targets, and the reflections, generally mixed with signals of other sources, can be received by at least two receiving antennas 4. The signals received by the receiving antennas 4 are called receiver signals. The receiver signals are first amplified with an amplifier 6 and mixed in the mixer 5 with the signal at the output of the oscillator 2 such that wavelets develop, which show a frequency portion in the basic band. From the wavelets, which are filtered via a band pass filter 7, the distance and the relative speed of a target, which cause a reflection, are determined in the control means 1, particularly from a Doppler shift and phasing, as disclosed in the patent publication DE 100 50 278 B4. For this purpose, the wavelets are digitized at the input of the control means via an analog-digital converter 8.
Based on delay times at the two receiver antennas 4 the incident angle of the reflections can be determined. The information gathered this way and additional information can then be forwarded to another processing step.
Between the cycles in which during operation of an above-mentioned radar device upchirps, downchirps, or Doppler chirps are transmitted, at certain intervals so-called calibration cycles are inserted. The calibration cycles have essentially two objectives:                the compensation of a frequency drift of the 24 GHz-mixer: Generally, a voltage controlled oscillator (VCO) is used, in which a frequency drift may occur, primarily caused by the abrupt change of temperature during operation, but also by other effects, such as for example load pulling or aging. The compensation of the frequency drift is required in order to avoid by all means that any respectively predetermined statutory frequency band limits are exceeded (cut-off frequency). The compensation occurs in every calibration cycle by an adjustment of the range of settings of the voltage controlled oscillator and/or a value respectively predetermined by the control means, by which the voltage controlled oscillator is adjusted.        The compensation of the non-linearity of the characteristic of the voltage controlled oscillator: The LFMSK-transmission process provides for a nested transmission of the three signal portions A, B, and C according to FIG. 2, with each of the three signal portions showing a frequency progression with equidistant frequency levels. Upholding these constant distances between two neighboring frequencies of a signal portion is of eminent importance for the target directive. Based on the non-linearity of the characteristic of the voltage controlled oscillator (frequency via adjusted voltage) for the setting of equidistant frequency levels here non-equidistant adjustment voltages develop and/or non-equidistant values predetermined by the control means for the digital-analog converter. They must be newly calculated in every calibration cycle for each of the transmission frequencies to be adjusted, because the progression of the characteristic of the voltage controlled oscillator depends on many factors, such as temperature, load pulling or aging, and thus permanently changes during operation.        
The calibration of the voltage controlled oscillator occurs in the radar device of the applicant via a calibration signal generated in the control means 1, which is supplied to a mixer 5 instead of a receiver signal. The calibration signal is mixed with the signal generated by the voltage controlled oscillator 2 in the mixer 5. The mixed signal is then forwarded via the receiver channel to the control means 1 and used for calibrating the voltage controlled oscillator 2.
In addition to a calibration, with the radar device of the applicant shown in FIG. 1 the detection of a malfunction of a receiver channel of the radar devices is also possible. The detection of a malfunction is possible during operation, i.e. during the upchirps, the downchirps, or the Doppler chirps.
The high frequency signal received by the receiver antenna 4, which shows the reflections of the transmitted signal of objects to be detected in the environment of the radar device, is supplied via an amplifier 6 to the mixer 5. Here, by the (coherent) mixing with the signal generated by the voltage controlled oscillator 2 a basic band signal develops, with the progression of its amplitude in an ideal mixer exclusively being determined by the phasing of the receiver signal in reference to the signal generated by the voltage controlled oscillator. However, the output signal of a real mixer 5 includes, in addition to the above-mentioned mixing product, also a so-called parasitic portion, which is also called mixer bias.
This mixer bias is dependent on the absolute frequency of the signal generated by the voltage controlled oscillator at 24 GHz. In the frequency band used showing 100 MHz and/or 200 MHz, approximately a linear dependency can be assumed. The precise parameters of this dependency are different, though, from one mixer 5 to the next mixer 5 due to parts and soldering tolerances of the high-frequency components, particularly the mixer diodes, being arbitrary and variable both quantitatively as well as qualitatively.
Due to the fact that the dynamic of the mixer bias in the output signals of the mixers of the receiver channels can be considerable without additional measures and might lead to a relevant worsening of the target detection, the analog filtering following in the mixer is not only designed as a low pass filter for limiting the noise but also as a band pass filter 7, in order to dampen the low-frequency mixer bias in the receiver signal. However, the damped portion of the mixer bias is present in the receiver signals and in this form it is the basis for detecting a channel failure. This detection occurs digitally, because the output signal of the band pass filter is already converted from analog to digital by an analog-digital converter 8 for the other signal processing means at a resolution of 12 Bit.
The option known to the applicant of detecting a channel failure comprises here to estimate the mixer bias, which due to its low frequency changes only very slowly over time, using an adaptive algorithm. When this algorithm yields an expected signal portion, it can be assumed that the portion of the receiver channel operates without malfunctions from the mixer 5 to the analog-digital converter 8. However, if no expected signal portions are provided, here a malfunction of a channel is very probable, for example by an interrupted contact or a defective part between the mixer 5 and the analog-digital converter 8.
The detection of a channel failure requires storage and computing capacities of the control means 1, which must be provided in addition to the storage and computing capacities of the control means 1 for target detection because the detection of a channel failure occurs during target detection. This can be considered disadvantageous.
Further, the target detection in the frequency ranges of the mixer bias required a strong damping of the basic band signal in the receiver channels. This is counter productive for detecting a channel failure, because the mixer bias is here damped to such an extent that a functional receiver channel can erroneously be detected as being defective.